Digital subscriber line noise mitigation techniques, and applications thereof

ABSTRACT

The present invention provides digital subscriber line noise mitigation techniques, and applications thereof. In an embodiment, the present invention provides a toolbox of methods and techniques for mitigating the effects of noise in xDSL systems. These methods and techniques are controllable and locatable at one or both ends of a DSL communication link (e.g., within a central office transceiver unit or a remote transceiver unit). These novel methods and techniques include: (1) per tone noise margin modification, (2) data framer constraints modification, (3) improved noise measurements; (4) more robust on-line reconfiguration processes, (5) worst case noise monitoring, (6) induced bit rate limitations, and (7) distortion noise mitigation. These methods and techniques are particularly useful for mitigating the effects of time-varying noise and impulse noise.

CROSS REFERENCE TO RELATED APPLICATION

This application claims the benefit of U.S. Provisional PatentApplication No. 60/783,071, filed on Mar. 17, 2006, which isincorporated herein by reference in its entirety.

FIELD OF THE INVENTION

The present invention generally relates to broadband communications.More particularly, it relates to digital subscriber line noisemitigation techniques, and applications thereof.

BACKGROUND OF THE INVENTION

Digital subscriber line (DSL) technology is used to transform anordinary telephone line (e.g., copper wire twisted-pair) into abroadband communication link. It works by sending signals over thetelephone line in previously unused high frequencies.

Over the years, DSL technology has evolved into a family of specific,standardized implementations. These various implementations offer avariety of transmission speeds and transmission distances. It is commonto refer to the various DSL implementations that have evolved over theyears collectively as xDSL.

Each of the various xDSL implementations typically employs eithercarrierless amplitude and phase (CAP) modulation or discrete multi-tone(DMT) modulation. CAP modulation is closely related to quadratureamplitude modulation (QAM). CAP modulation produces the same form ofsignal as QAM without requiring in-phase and quadrature components ofthe carrier to first be generated. DMT modulation is a modulation methodin which the available bandwidth of a communication channel is dividedinto numerous subcarriers or tones. Each tone of a DMT communicationsystem is capable of acting as a communications sub-channel that carriesinformation between a transmitter and a receiver.

A number of factors determine the performance of the various xDSLimplementations. For example, the performance of any of the xDSLimplementations is highly dependent on the local loop length (e.g., thelength of a twisted-pair circuit between a central office and acustomer) and the local loop condition. The local loop condition isaffected by several factors such as, for example, line noise. Line noisemay corrupt data-bearing signals as the signals travel along the line.As a result, the transmitted data-bearing signals may be decodederroneously by a receiver because of this signal corruption.

In the case of DMT modulation, for example, the number of data bits orthe amount of information that a tone carries may vary from tone totone, and it depends on the relative power of the data-bearing signalcompared to the power of the corrupting signal on that particular tone.A measure of the quality of a signal carried by a tone is the ratio ofthe received signal strength (power) over the noise strength in thefrequency range of operation, or the Signal-to-Noise Ratio (SNR). HighSNR results in high signal quality being carried by a tone. Anothermeasure of signal quality is bit error ratio (BER) for a given tone.

In order to account for potential interference on the telephone line andto guarantee a reliable communication between the transmitter andreceiver, each tone is typically designed to carry a limited number ofdata bits per unit time based on the tone's SNR using a bit-loadingalgorithm. The number of bits that a specific tone may carry decreasesas the relative strength of the corrupting signal increases, that iswhen the SNR is low. Thus, the SNR of a tone may be used to determinehow much data should be transmitted by the tone.

It is often assumed that the corrupting signal is a stationary additiverandom source with Gaussian distribution and white spectrum. With thisassumption, the number of data bits that each tone can carry relatesdirectly to the SNR. However, this assumption may not be true in manypractical cases and there are various sources of interference that donot have a white, Gaussian distribution. Impulse noise is one such noisesource. Time-varying noise is also an effect that makes the stationarityassumption impractical.

Bit-loading algorithms, which are methods to determine the number ofbits transmitted per tone, are usually designed based on the assumptionof stationary additive, white, Gaussian noise. With such algorithms, theeffects of impulse noise and time-varying noise are misestimatedresulting in an excessive rate of error. Furthermore, channel estimationprocedures can be designed to optimize performance in the presence ofstationary, additive, white, Gaussian noise, but are often poor atestimating impulse noise and time-varying noise. Consequently, DSL modemtraining procedures leave the modem receivers susceptible to impulsenoise and time-varying noise.

What are needed are new DSL noise mitigation techniques that overcomethe deficiencies noted above.

BRIEF SUMMARY OF THE INVENTION

The present invention provides digital subscriber line noise mitigationtechniques, and applications thereof. In an embodiment, the presentinvention provides a toolbox of methods and techniques for mitigatingthe effects of noise in xDSL systems. These methods and techniques arecontrollable and locatable at one or both ends of a DSL communicationlink (e.g., within a central office transceiver unit or a remotetransceiver unit). These novel methods and techniques include: (1) pertone noise margin modification, (2) data framer constraintsmodification, (3) improved noise measurements; (4) more robust on-linereconfiguration processes, (5) worst case noise monitoring, (6) inducedbit rate limitations, and (7) distortion noise mitigation. These methodsand techniques are particularly useful for mitigating the effects oftime-varying noise and impulse noise.

Further embodiments, features, and advantages of the present invention,as well as the structure and operation of the various embodiments of thepresent invention, are described in detail below with reference to theaccompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS/FIGURES

The accompanying drawings, which are incorporated herein and form a partof the specification, illustrate the present invention and, togetherwith the description, further serve to explain the principles of theinvention and to enable a person skilled in the pertinent art to makeand use the invention.

FIG. 1 is a diagram illustrating an example digital subscriber linearchitecture.

FIG. 2 is a diagram of a central office transceiver unit and a remotetransceiver unit according to an embodiment of the present invention.

FIG. 3 is a diagram of a discrete multi-tone transceiver according to anembodiment of the present invention.

FIG. 4 is a diagram of a transmitter data path for the transceiver ofFIG. 3.

FIG. 5 is a diagram of a receiver data path for the transceiver of FIG.3.

The present invention is described with reference to the accompanyingdrawings. The drawing in which an element first appears is typicallyindicated by the leftmost digit or digits in the corresponding referencenumber.

DETAILED DESCRIPTION OF THE INVENTION

The present invention provides digital subscriber line noise mitigationtechniques, and applications thereof. In the detailed description of theinvention herein, references to “one embodiment”, “an embodiment”, “anexample embodiment”, etc., indicate that the embodiment described mayinclude a particular feature, structure, or characteristic, but everyembodiment may not necessarily include the particular feature,structure, or characteristic. Moreover, such phrases are not necessarilyreferring to the same embodiment. Further, when a particular feature,structure, or characteristic is described in connection with anembodiment, it is submitted that it is within the knowledge of oneskilled in the art to effect such feature, structure, or characteristicin connection with other embodiments whether or not explicitlydescribed.

FIG. 1 is a diagram illustrating an example digital subscriber line(DSL) architecture 100. Architecture 100 includes a customer 102connected to a central office 104 by a local loop 106. Local loop 106 isused to provide DSL service to customer 102.

As shown in FIG. 1, a baseband service terminal (such as a POTS or anISDN terminal) 108 can connect to local loop 106 using network interfacedevice (NID) 110. A computer 112 can connect to local loop 106 using NID110 and a local loop remote end modem or transceiver unit-remote (TU-R)114. A signal splitter (not shown) within NID 110 is used to pass lowfrequency baseband signals to terminal 108 and high frequency datasignals to TU-R 114. The signal splitter includes filters designed todiplex signals onto local loop 106, and they have stopband impedancecharacteristics that minimize the effect of changing baseband service108 signal characteristics. This allows customer 102 to use local loop106 as a communication link for both baseband service and for broadbanddata service.

Central office 104 includes a signal splitter 116, a DSL accessmultiplexer (DSLAM) 120, and a switch/router 124. Signal splitter 116passes low frequency telephone signals received from telephone 108, overlocal loop 106, to the baseband service network (PSTN or ISDN) 118. Highfrequency data signals received over local loop 106 are passed to DSLAM120. DSLAM 120 includes a plurality of central office modems ortransceiver units-central office (TU-C) 122 (i.e., one TU-C for eachcustomer 102) arranged in a bank configuration. From DSLAM 120, the highfrequency data signals are sent to switch/router 124 and transmittedover internet 126. Signals from ISDN/PSTN 118 and internet 126 tocustomer 102 are sent by central office 104 to customer 102 over localloop 106.

For residential and commercial customers 102 with a need for broadbanddata access, but who do not typically send large data streams, the DSLservice provided is typically asymmetric DSL (ADSL) service. Thisservice is so named because the data rate sent to customer 102 is muchgreater than the data rate sent by customer 102. Other customers 102,who need to send large data streams are typically provided a symmetricalxDSL service such as VDSL or SHDSL.

For the example of ADSL, the low frequency spectrum allocated for use bytelephone 108 is from near DC to approximately 4 kHz. A frequency guardband is placed between this spectrum and the high frequency dataspectrum to help avoid interference. The high frequency data spectrumstarts above the telephone/ISDN 108 band and extends up to approximately1.1 MHz. In embodiments, the lower part of the high frequency spectrum(i.e., the upstream spectrum) is used to send data from customer 102 tocentral office 104. The upper part of the high frequency spectrum (i.e.,the downstream spectrum) is used to send data from central office 104 tocustomer 102.

FIG. 2 is a more detailed diagram of TU-C 122 and TU-R 114 according toan embodiment of the present invention. As shown in FIG. 2, TU-C 122includes a transceiver 200 that includes a noise mitigation engine 202and a noise mitigation controller 204. Noise mitigation engine 202communicates with noise mitigation controller 204 over communicationlink 216. TU-R 114 includes a transceiver 206 that includes a noisemitigation engine 208 and a noise mitigation controller 210. Noisemitigation engine 210 communicates with noise mitigation controller 208over communication link 218. Noise mitigation controller 204 and noisemitigation controller 210 communicate with each other over communicationlink 212. Transceiver 200 communicates with transceiver 206 over DSLcommunication link 214. In an embodiment, communication link 212 is areserved communication sub-channel of DSL communication link 214.

Noise mitigation engines 202 and 208 operate under the control of noisemitigation controllers 204 and 210 to improve the quality of DSL serviceprovided between transceivers 200 and 208. The noise mitigation enginesare necessary in order to provide a high quality of service becausethere are several types of interference that can potentially limit theperformance of xDSL systems. These include, for example, additive whiteGaussian noise interference, crosstalk interference, impulse noiseinterference, and radio noise interference. Additive white Gaussiannoise is thermal noise, and it can cause symbol errors to occur at anxDSL receiver when noise pushes the received signal beyond a decisionboundary. Crosstalk occurs because telephone cables contain many bundledwires (e.g., twisted pairs), each used by a different customer 102. Thebundled wires radiate electromagnetically and can induce currents inother wires in the cable. Impulse noise is interference that is short induration but relatively large in magnitude. It can be caused, forexample, by power electronic devices, by a power surge that results whena device having a motor is started, or by lightening. Radio noise isinterference due to a wireless source. The copper phone lines act asantennae and pick up this interference.

There are two types of crosstalk interference. Near-end crosstalkinterference occurs when a transmitter interferes with a receiverlocated on the same end of the cable. Far-end crosstalk interferenceoccurs when the transmitter interferes with a receiver on the oppositeend of the cable. The effect of near-end crosstalk interference isdifferent than far-end crosstalk interference because far-end crosstalkmust travel the entire length of the cable and is attenuated when itreaches the receiver, the net effect of upstream and downstream beingalso determined by the power spectral density of cross-talk signals andcoupling factor being frequency and cable dependent.

In addition to the above, bridged taps and line splices are also asource of signal perturbation that can potentially limit the performanceof xDSL systems. A bridged tap is a section of wire connected to a localloop at one end and unterminated at the other end. When a transmittedsignal arrives at a bridged tap, the signal divides. Part of the energycontinues to the receiver and the rest of the energy reflects off of theunterminated end. This reflection causes delayed versions of the signalto arrive at the receiver, and these reflections distort the receivedsignal causing both intersymbol interference and intrasymbolinterference. Line splices have a similar effect.

A more detail description of the features and operation of TU-C 122 andTU-R 114 are provided below reference to FIGS. 3-5.

FIG. 3 is a diagram of a transceiver 300 according to one embodiment ofthe present invention. Transceiver 300 is a discrete multiple-tone (DMT)transceiver. As noted above, two types of modulation are typically usedby xDSL systems: carrierless amplitude and phase (CAP) modulation anddiscrete DMT modulation. CAP modulation is closely related to quadratureamplitude modulation (QAM) and produces the same form of signal as QAMwithout requiring in-phase and quadrature components of the carrier tofirst be generated. DMT modulation transmits data on multiplesubcarriers in a manner similar to the orthogonal frequency divisionmultiplexing (OFDM) technique that is used in many wirelessapplications.

Transceiver 300 includes both a transmitter data path, a receiver datapath and a noise mitigation engine 322. The transmitter data pathincludes a serial-to-parallel input data buffer 302, a transmit tonedata processor 304, an inverse fast Fourier transformer (IFFT) 306, adigital-to-analog converter (DAC) 308, and a filter 310. The receiverdata path includes a filter 312, an analog-to-digital converter (ADC)314, a fast Fourier transformer (FFT) 316, a receive tone data processor318, and a parallel-to-serial output data buffer 320. Noise mitigationengine 322 includes a tone data table 323.

The transmitter data path of transceiver 300 takes in N data symbols inparallel and transmits the symbols on N subcarriers or tones (i.e.,tones T₁ through T_(n)). The data rate on each tone is 1/N the originaldata rate. Reducing the data rate results in a DMT symbol period that isN times as long as the original symbol period. Increasing the symbolperiod can make the symbol longer than the channel response. This isused to mitigate the effects of intersymbol interference, or moregenerally the channel dispersion.

In addition to the above, transceiver 300 can perform dynamic bitallocation to make efficient use of the available channel capacity. Thistechnique enables transceiver 300 to vary the number of bits per symbolfor each tone based on the tone's signal-to-noise ratio (SNR). Toneswith a low SNR transmit binary phase-shift keying (BPSK) or quadraturePSK (QPSK) because they are robust modulation formats. If the tone's SNRis very low, that tone will not be used to transmit data at all. Toneswith a higher SNR transmit higher-order quadrature amplitude modulation(QAM) in order to achieve an increased throughput.

Like many other digital communication systems, xDSL systems employerror-control coding to help mitigate the effect of noise such as, forexample, additive white Gaussian noise and radio noise. This coding isperformed by transmit tone data processor 304. The coding addsredundancy to the transmitted signal, which is used by a receiver todetect and correct errors.

In one embodiment, transceiver 300 employees three layers of coding. Theinnermost code is a convolutional code. Convolutional codes get theirname because the encoding process can be viewed as the convolution ofthe message with the code's impulse response. The Viterbi algorithm isused at the receiver to decode the received sequence. Convolutionalcodes are good at correcting random errors. However, the nature of thedecoding algorithm is such that the decoder can cause burst errors tooccur if errors are made during the decoding process.

In an embodiment, a Reed-Solomon block code is used on top of theconvolutional code. Reed-Solomon codes are good at detecting andcorrecting burst errors, such as those generated by the Viterbi decoder.Reed-Solomon code-word lengths of up to 255 bytes with the addition ofup to 16 parity bytes can be used for each code word.

In an embodiment, the outermost code is a cyclic redundancy check (CRC)code. The CRC can detect errors, but it cannot correct them. The CRCcode is used as a top-level error-detection mechanism in order to detectany errors that remain after Viterbi and Reed-Solomon decoding.

Transceiver 300 preferably employs interleaving in combination withcoding to correct errors caused by impulse noise. The interleavingprocess rearranges data so that those samples that were locatedcontiguously in time are spaced far apart. Impulse noise can cause aburst of errors that is hard for conventional decoders to correct. Theuse of interleaving combined with coding spreads out these errors intime to improve decoding performance.

Following coding by transmit tone data processor 304, the coded symbolsare forwarded to IFFT 306. IFFT 306 generates orthogonal subcarriers.The data symbols are treated as being in the frequency domain and act ascomplex weights for the basis functions (orthogonal sinusoids atdifferent frequencies) of the transform performed by IFFT 306. IFFT 306converts the data symbols into a time-domain (e.g., sum of sinusoids)signal. The block of IFFT output samples is referred to as a DMT symbol.This time-domain signal is converted to an analogy signal by DAC 308,filtered by filter 310, and transmitted across DSL communication link214.

In an embodiment, a 2N-point inverse fast Fourier transform is used togenerate the DMT symbols, and the N negative-frequency inverse fastFourier transform bins are the complex conjugate of the Npositive-frequency bins. This symmetric spectrum results in a realtime-domain signal. The DMT signal is centered at DC with thesubcarriers around DC zeroed out (i.e., not used) to create a hole inthe DMT spectrum in order to make room for the low frequency telephonesignal.

DMT supports inclusion of cyclic prefix and postfixes. A cyclic prefixor postfix is a block of samples with a length L that is a replica ofthe last or first L samples of the DMT symbol. The prefix is transmittedfirst or last, followed or preceded by the 2N samples of the DMT symbol.The length L is chosen so that it will be longer than the length of thechannel response, or in conjunction with the postfix length in a waythat mimizes the effect of the channel dispersion. The cyclic prefixcontains redundant information. However, the DMT receiver makes use ofthe presence of the prefix in order to mitigate the effects of the DSLcommunication link 214.

FIG. 4 is a more detailed diagram of the transmitter data path oftransceiver 300 according to an embodiment of the present invention. Inthe embodiment shown in FIG. 4, the transceiver data path includesserial-to-parallel input data buffer 302, transmit tone data processor304, IFFT 306, a cyclic prefix inserter 400, DAC 308, an amplifier 402,and filter 310.

Referring to FIG. 3 again, input signals received by transceiver 300 arefiltered by filter 312 and converted to digital signals by ADC 314. FFT316 then transforms the digital time domain signals to frequency domainsignals. Receive tone data processor 318 corrects transmission errorsand reverses the coding process of transmit tone data processor 304 toform data that is written to parallel-to-serial output data buffer 310.

FIG. 5 is a more detailed diagram of the receiver data path oftransceiver 300 according to an embodiment of the present invention. Inthe embodiment shown in FIG. 5, the receiver data path includes filter312, an amplifier 500, ADC 314, a time domain equilizer 502, a cyclicprefix remover 504, FFT 316, an frequency domain equalizer 506, receivetone data processor 318, and a parallel-to-serial output data buffer310.

Multicarrier systems such as the example system described above containmany simultaneous signals. If each signal's peak amplitude isrepresented by X, and all signals simultaneously reach peak signallevel, the resulting level would be X*20 log(N), where N is the numberof signals. The signals in such a system have a statistical nature,however, and can be treated as uncorrelated. Thus, the possible peakamplitude may be large, but the probability of this level occurring islow.

A peak-to-average ratio (R_(p)) is used to define the ratio between asignal's peak level and its average level over time. Most multicarriersystems use a modified definition for R_(p) that is based on thestatistical likelihood of exceeding a certain peak level (such as theprobability of clipping in the DAC output). The R_(p) value partiallydetermines the operating parameters for DAC 308 and ADC 314.

Important parameters relating to the example DSL system described aboveare the R_(p) factor, the number of bits per subcarrier or tone, and thesignal-to-noise ratio (SNR). ADC 314 must take into account all of theseparameters, plus additional bits of resolution for input noise (i.e.,the receiver SNR is typically lower than transmitter SNR) and forpossible echo energy. Typically one to two extra bits of resolutionshould be used in ADC 314.

Referring back to FIG. 2, during operation, when TU-R 114 is firstconnected to DSL communication link 214, TU-R 114 and TU-C 122 performan initialization process. This process typically consists of four majorphases. The first phase is a handshake phase. The second phase is atransceiver training phase. The third phase is a channel analysis phase.The fourth phase is a data exchange or showtime phase.

The handshake phase is used to determine the nature and capabilities ofTU-R 114 and TU-C 122 and to indicate which protocol is to be used forthe remainder of the initialization phase. The signaling method used forthe handshake interchange is preferably robust in case the channelcharacteristics of DSL communication link 214 are poor. Bi-phase shiftkeying (BPSK) modulation is preferably used to modulate multiplesingle-tone subcarriers that each carry the same data. The tones usedare selected based on impairments likely to be present. The handshakecan have many variations, but TU-R 114 and TU-C 122 should exchange amessage that contains information about the their type and a number ofrelated subparameters such as the frequency range and number of DMTsubcarriers supported.

The second phase of initialization is transceiver training. In thisphase, transceivers 200 and 206 acquire the DMT symbol stream, adjusttheir receiver gain, perform symbol timing recovery, and train anyequalizers they may contain. There is an optional echo cancellationtraining step that can also be performed during this phase.

The transmitter power of transceivers 200 and 206 is initially set to apredetermined level during transceiver training to allow a preliminaryestimate of loop attenuation by transceivers 200 and 206. The receivedupstream power spectral density (PSD) level is reported back to the TU-R114, for example, via communications link 212 to allow limited powerlevel adjustment, if needed, to meet spectral mask requirements. Thetraining phase is conducted with all available upstream and downstreamsubcarriers modulated using.

In the third phase, transceivers 200 and 206 exchange capabilityinformation and perform detailed channel characterization. For example,TU-C 122 may specify the minimum SNR margin for the DSL system andwhether it can support functions such as trellis coding and echocancellation. Similar information is exchanged by TU-R 114.

During this third phase, transceivers 200 and 206 both attempt tomeasure specific channel characteristics such as unusable subcarriers,loop attenuation on a per subcarrier basis, SNRs, and any other channelimpairments that would affect the potential transmitted bit rates.

In an embodiment where multiple orthogonal subcarriers are used, eachsubcarrier can be assigned a modulation format (e.g., number of bits persubcarrier) and relative gain independently. TU-C 122 assigns bits andgains for the downstream spectrum while TU-R assigns bits and gains forthe upstream spectrum. The last part of the exchange phase is asynchronized transition from the BPSK and QPSK used duringinitialization to the QAM modulation assigned during the data exchangephase. At the conclusion of the initialization steps, the system isready to pass higher-layer traffic.

As described in detail below, it is a feature of the present inventionthat it provides a toolbox of methods and techniques for mitigating theeffects of noise in xDSL systems. These methods and techniques arecontrollable and locatable at one or both ends of DSL communication link214 (e.g., within TU-R 114 and TU-C 122). These novel methods andtechniques include: (1) per tone noise margin modification, (2) framerconstraints modification, (3) improved noise measurements; (4) morerobust on-line reconfiguration processes, (5) worst case mean noisemonitoring, (6) induced bit rate limitations, and (7) distortion noisemitigation. The source(s) of information used with these novel noisemitigation methods and techniques come from noise mitigation engine 202or noise mitigation engine 208, which can each monitor noiseimpulsivity, from average power values, from locally provided settings(e.g., set by a modem vendor), or from noise mitigation controller 204or noise mitigation controller 210.

In the per tone noise margin modification method of the presentinvention, TU-R 114 and TU-C 122 each measure the impulsive noise natureof the noise (e.g., using noise mitigation engines 202 and 208) duringat least a portion of the initialization process described above using amedley signal on a per bin basis. This method then applies a margincorrection to mitigate the effect of the noise impulsivity. In anembodiment, TU-R 114 and/or TU-C 122 may measure the impulsivity of thenoise using signals other than a medley signal such as, for example, areverbs signal or a quiet signal. Optionally, either TU-R 114 and/orTU-C 122 may disable (or not disable) this behavior independently.

In an embodiment, TU-R 114 uses a per-tone additional per tone marginpattern that is either programmed locally or communicated to TU-R 114 byTU-C 122. This pattern data is stored, for example, in a tone data tablesuch as tone data table 323. Using this data, TU-R 114 is able tocompute an effective tone dependent target margin. A predefined (empty)command can be used to disable the behavior at TU-R. Employing per tonemargin control is preferable to using a target margin that is given foran entire frequency band, which cannot take into account frequencylocalization of some noises.

In an embodiment, there are multiple sources for the additive per tonemargins stored in tone data table 323. A first source is an externallydefined table, defaulting to a zero-filled table. An example applicationin which this is useful is where a circuit board supply has identifiedfrequency domain noise issues such as, for example, distortion duringproduction. In this example case, per tone corrections can be determinedand stored in the external table. A second source is values obtainedduring the last data exchange phase or showtime (e.g., values stored ina retrain controller object). The use of this source may be controlledexplicitly, for example, in the physical layer by an enable/disablecontrol bit. The way this additional target per tone target margin isstored can be implementation specific, based on the results of anautonomous impulsivity measurement. A third source is an autonomousdetermination of the additional margin to be used, done during trainingin the initialization process. This can be based on the average noise,the peak noise, and the probability that the excess threshold of thenoise distribution sigma (e.g., TrainingPlnThresholdEquivalentSigma) wasexceeded. The quantities are measured by the same impulsivity primitivesused in showtime. In an embodiment, the use of this source may also becontrolled by a control bit. As will be understood by persons skilled inthe relevant art(s) given the description herein, other sources are alsopossible, and thus the present invention is not to be limited to justthese example sources.

In an embodiment, table 323 is built using two additional impulsivitymeasurements made during training. Table 1 below provides examplepseudo-code that illustrates one way to due this. In the examplepseudo-code, threshold, classes and additive margin level may beimplementation specific and/or specific to TU-R 114.

TABLE 1 If enabled, If Prob-ExcessThresholdPeakImpulseNosie(tone)>Threshold_do_add_margin or forced, additiveTargetMargin (tone) +=addMarginPeakImpulseNoise [Class [peak_noise (tone) ] ] elseadditiveTargetMargin (tone) += addMarginPeakImpulseNoise [Class[peak_noise (tone) ] ]

Another source for the additive per tone margins data is from a tableresiding within TU-C 122. In an embodiment, a request is made to TU-C122 by TU-R 114 for the data. The request can be made using an interfacefield.

In an embodiment, the default behavior for TU-R 114 is that it uses themaximum of the available sources as additional margin. In an embodiment,TU-C 122 may disable any of these sources (e.g., using an interfacefield) in order to make sure that data from a table within TU-C 122,which TU-C 122 provides to TU-R 114, is used. A vendor of TU-R 114 canprovide a user with access to the control bits through a dedicatedcontrol interface (e.g., a GUI or a CLI). This will indicate to TU-R 114control logic that it to use locally enabled impulsivity measurementsfrom either showtime or training to compute per tone additional margin.

In an embodiment, TU-C can also apply the same principle described hereto its own margin allocation.

In an embodiment, one or more control bit(s) are provided that can beused to prevent TU-R 114 or TU-C 122 from using any of the noisemitigation methods and techniques described herein. In anotherembodiment, the bit(s) are used to enable selected noise mitigationmethods and techniques while disabling other.

In the framer constraints modification method of the present invention,TU-R 114 and/or TU-C 122 measure broadband impulsivity characteristicsduring the initialization process and derive additional constrain(s) forthe receiver framer. In an embodiment, the addition constrain(s) relateto increased minimum impulse noise protection (INP), lower maximumdelay, and control of the use of trellis coding. These constrain(s) area function of the impulsivity measurements. This method provides animprovement for all impulse noise processes, whether being repetitive ornot. Those that are repetitive with a high repetition frequency arecaptured using a medley signal.

In an embodiment, TU-R 114 use peak line noise information measuredduring the initialization process for framing parameter improvements.This induces a behavior such that TU-R 114 takes into account theimpulsivity of the noise explicitly in the framer algorithm, giving thepreference to impulse noise resistance to other aspects such as bitrate. Both TU-C 122 and TU-R 114 may negotiate longer than normal medleysignal durations. In an embodiment, this longer duration is configurablein TU-C 122 (e.g., using a human machine interface) and is transmittedto TU-R 114 for use (e.g., using an interface field).

In an embodiment, TU-R 114 noise mitigation engine 208 is capable ofderiving additional framer constrains from a previous showtime or fromimpulsivity measurements performed during training. Noise mitigationengine 208 obtains broadband histograms of both inter-arrival andimpulse duration during training or during a previous showtime, forcontrolling TU-R 114. The duration bins preferably have a one symbolresolution and go up to 32. The inter-arrival bins preferably correspondto a period corresponding to integer values of milliseconds. The margindecrease levels are set to specified levels (e.g.,PerToneMarginDecreaseLevelPeakImpulseNoiseTraining) for the per-tonethreshold and for the broadband threshold (e.g.,Broadband/MarginDecreaseLevelPeakImpulseNoiseTraining).

In an embodiment, TU-R 114 noise mitigation controller 210 has twobehaviors. The first behavior is to use locally defined TU-R 114 valuesfor the maximum interleaving delay, the minimum impulse noise protectionvalue, and the trellis coding. The second behavior uses the broadbandhistograms obtained during the last showtime and stored in a retraincontroller object. This behavior is based on a measurement ofimpulsivity histograms of broadband noise events measuring thedistribution of the broadband pulse durations and the broadband pulseinter-arrival times. The impulsivity showtime functions are preferablyreused during medley.

In an embodiment, if the above method is not disabled by TU-C 122 (e.g.,using an interfacing field), the operation of TU-R 114 is effected basedon inter-arrival statistics. For example, let P1 be the probability ofobserving inter-arrival times corresponding to values of the delay inmilliseconds lower than a negotiated maximum delay. If P1 is higher thana threshold percent (e.g., AggregateInterArrival/DelayChangeThreshold),the framing algorithm uses a value of the maximum delay such that athreshold percent (e.g.,IADelayChangeFraction*AggregateInterArrival/DelayChangeThreshold) of thepulses observed have an inter-arrival time between this value and anegotiated value of the maximum delay. All other inter-arrival timeobservations are below the chosen maximum delay.

The impulse noise protection minimum value preferably used is theduration for which a threshold percent (e.g.,AggregateDurationThreshold) of the broadband pulse events observed arestrictly shorter. This value is used in preference to the negotiatedminimum impulse noise protection value, but only if the minimumdownstream bit rate is lower than the maximum feasible rate at thatimpulse noise protection value, and if this feature is not disabled byTU-C 122 to avoid a configuration not possible. An impulse noiseprotection value resolution of one is implemented in the framingalgorithm.

In an embodiment, if the number of detected pulses is above apredetermined percent (e.g., AggregatePulseRateForTrellisOff), TU-R 114request no down stream trellis coding. In an embodiment, downstreamtrellis coding can be configured through an interface field by TU-C 122directly.

In the improved measurements method of the present invention, both TU-R114 and TU-C 122 may negotiate a longer medley signal duration forimproved robustness of the impulse noise measurement. Based on aprevious showtime, both entities may require a longer medley signal,within limits, to confirm that the detected impulsivity measurements arestill present and confirm that it may be desirable to apply ad-hoc noisemitigation methods. This extension applies for those noises that arerepetitive enough to be measurable during an extended medley signal. Inan embodiment, these measurements are done during a loop diagnosticsmode where a long medley signal is sent and where additional messages(e.g., R-LD-x and C-LD-x in an ADSL2 embodiment) are exchanged tocommunicate measurement results.

In an embodiment of the present invention, TU-R 114 adjusts its behaviorbase on worst-case line noise in showtime. This feature permits TU-R 114to take into account the impulsivity and the time variance of the noiseexplicitly during showtime, giving preference to impulse noiseresistance over other aspects such as bit rate. This induces the on-linereconfiguration mechanisms to keep higher margin on those tones wherehigh impulsivity level is detected.

In an embodiment, TU-R 114 performs worst-case noise monitoring. In thisembodiment, TU-R 114 maintains a worst case per tone noise table. Thesenoise values are not a peak impulse noise value, but an average value ofthe noise, as measured during the SNR showtime estimation. The initialvalue of the worst case noise may be programmed into TU-R 114, in orderto prevent the physical layer program code from artificially limitingthe measured SNR and the bit allocation. This is particularly useful toavoid the influence of frequency dependent board noise when componentsintroducing distortion-originated impulses are used, or to complementthe line-length dependency of transmit noise protection from TU-C 122.

In an embodiment, the difference between worst case line noise andinitial quite line noise is preferably averaged over the usable band tocompute the allowable average target margin lowering. The update of theworst case noise is enabled or disabled through a control bit. If theupdate is disabled, the initial value is permanently used as the minimalvalue of the noise power. TU-C 112 is able to reset the worst case noisemonitoring during showtime using an embedded operations channel commandextension that may be implementation specific. In such a case, the TU-C122 transmitted table is max-ed to the initial value. The update canalso be controlled by TU-C 122.

In an embodiment, the bit swap algorithm takes into account the worstcase noise on each tone to perform new bit and gain allocations. If theaverage target margin lowering is less than the threshold communicatedby TU-C 122 (the default value being target margin−minmargin+AdditionalTargetMarginRelaxationForBitswap dB), the bit swap isperformed. If the average target margin lowering level is higher thanthe communicated threshold, TU-R 114 induces a retrain and uses worstcase noise during the retrain in preference to the quite line noisemeasured during initialization. The amount of available average margincan be increased by TU-C by either increasing the target margin or byusing an interface field based additional target margin field.

In an embodiment, a seamless rate adaptation (SRA) algorithm takes intoaccount the worst case noise on each tone to perform new bit and gainallocations. If an increase of the target margin was set by TU-C 122,however, it is maintained when the seamless rate adaptation algorithm isperformed. This behavior can be controlled by TU-C 122 usingenable/disable commands.

As noted above, it is a feature of the present invention that itprovides more robust on-line reconfiguration processes. Theseimprovements include, for example, if repetitive impulse noise detectionis enabled, downstream full-band bit swap and seamless rate adaptationare limited to non-segmented requests, and providing overhead messagedoubled bit swap confirmations. These features lower the errored secondscount and lower the probability of further stability degradation, ifpeak line noise is detected.

In an embodiment, on-line reconfiguration is based on autonomous peakline noise detection. The principle of this feature is that whentriggered by an excessive error rate of a predetermined number oferrored seconds (e.g., PercentageOfEStoTriggerOLR) within apredetermined period of time (e.g.,AutonomousPeakImpulseNoiseTriggerMonitorError), an autonomous peak linenoise measurement is started. If a significant peak line noise level isfound, an equivalent target margin per tone is computed and is used in asubsequent on-line reconfiguration (e.g., either bit swaps or seamlessrate adaptation).

A bit swap is applied upon detection of a sync symbol phase inversion.In an environment where impulse noise is present, there is a significantprobability that a bit-swap acknowledge is missed by the receiving part,or that a bit-swap acknowledge is wrongly detected earlier than theeffective bit sway. This will led to errors being made until theeffective requested bit swap is made, and it will make local errormonitoring and the adaptive algorithm biased.

When TU-R 114 or TU-C 122 has detected impulse noise events exceedingpredetermined value (e.g., threshold1), it activates an overhead channelbased bit swap confirmation mechanism. The on-line reconfigurationrequests are limited to using non-segmented overhead messages. In anembodiment, this requires a bit swaps through the on-linereconfiguration value or-ed with 0×20. This means that a confirmation isexpected through the statement overhead channel. The responding party(TU-C 122 or TU-R 114) responds through the overhead channel with areason code 0×20 that the bit swap is accepted. If the requesting party(TU-R 114 or TU-C 122) receives this confirmation, it replies with thesame on-line reconfiguration message and applies the bit-swap atsuper-frame following the time-out. If the demanding party receives areason code different than 0×20, it reverses the applied bit-swap if async symbol inversion was detected. This feature is used if both TU-C122 and TU-R 114 indicate they support this feature (e.g., in aninterface frame).

In the worst case mean noise monitoring method of the present invention,TU-R 114 retains worst case noise during showtime to perform seamlessrate adaptation. This can be controlled by TU-C 122. TU-R 114 alsoretains worst case noise during showtime to perform bit swaps, which isalso controllable by TU-C 122. TU-C 122 is able to indicate to customerpremises equipment to take additional target margin to cope withstrongly varying noise. This feature of the present invention allows theon-line reconfiguration mechanisms to learn the envelope of the externalnoise variations, such as the crosstalk noise variation, and adapteither the bit rate if seamless rate adaptation is enabled, or the bitswaps. It is to be noted that the envelope of these noise variationswill extend over a time-frame of about 24-hours, and can deal with the7PM-crosstalk phenomenon, i.e. the fact that broadband usage peaks atsome hours of the day.

In an embodiment, TU-C 122 and TU-R 114 use worst case echo noise tolimit the effects of distortion. The principle of this mitigationtechnique is to use a high crest factor transmit signal when the echonoise proper is measured. This echo noise power is stored and comparedto the medley noise power used to compute the medley signal-to-noiseratio per bin. The echo noise power is an average measure, and not apeak measure. If the high crest echo noise is higher than the medleynoise, the former is used to compute the medley signal-to-noise ratio.This approach guarantees that the bit allocation is such that distortionnoise will not cause error events in the downstream spectrum. The valuesof the high crest echo noise are also compared to the initial worst casenoise values, and they are used if higher than them, on a per bin basis.As with other features of the present invention, this behavior can becontrolled by an enable/disable control kit.

In the induced bit rate limitations method of the present invention,TU-C 122, for example, adds transmit noise during the initializationprocess to induce a bit rate limitation at TU-R 114. This feature allowsTU-C 122 to inject noise at the transmit side and obtain an effect thatis totally controlled at the central office side.

In the distortion noise mitigation method of the present invention, TU-C122 or TU-R 114 takes into account the echo noise measured with a highcrest factor transmit signal as lower limit for medley noise.

While various embodiments of the present invention have been describedabove, it should be understood that they have been presented by way ofexample, and not limitation. It will be apparent to persons skilled inthe relevant art(s) that various changes can be made therein withoutdeparting from the scope of the invention. Furthermore, it should beappreciated that the detailed description of the present inventionprovided herein, and not the summary and abstract sections, is intendedto be used to interpret the claims. The summary and abstract sectionsmay set forth one or more but not all exemplary embodiments of thepresent invention as contemplated by the inventors.

1. A digital subscriber line transceiver, comprising: a noise mitigationengine configured to determine per-tone values based on a time-varyingnoise associated with each tone used to transmit data bits, and furtherbased on a medley signal wherein a duration of the medley signal isvaried depending on a noise impulsivity of the time-varying noise; and areceive tone data processor engine coupled to the noise mitigationengine, wherein the noise mitigation engine is further configured toapply a margin correction to each tone based on the per-tone values tomitigate the effect of the time-varying noise.
 2. The digital subscriberline transceiver of claim 1, wherein the per-tone values are determinedduring an initialization period.
 3. The digital subscriber linetransceiver of claim 1, wherein the per-tone values are determined afteran initialization period.
 4. The digital subscriber line transceiver ofclaim 1, wherein the time-varying noise is an impulse noise.
 5. Thedigital subscriber line transceiver of claim 1, wherein the per-tonevalues are used to adjust a minimum impulse noise protection value. 6.The digital subscriber line transceiver of claim 1, wherein the per-tonevalues are used to control trellis coding.
 7. A digital subscriber linetransceiver unit, comprising: a transceiver that includes a noisemitigation engine configured to determine and to store per-tone valuesbased on a time-varying noise associated with each tone used to transmitdata bits in a tone data table, and further based on a medley signalwherein a duration of the medley signal is varied depending on a noiseimpulsivity of the time-varying noise, wherein the noise mitigationengine is further configured to apply a margin correction to each tonebased on the per-tone values to mitigate the effect of the time-varyingnoise; and a noise mitigation controller coupled to the noise mitigationengine of the transceiver.
 8. The digital subscriber line transceiverunit of claim 7, wherein the per-tone values are determined during aninitialization period.
 9. The digital subscriber line transceiver ofclaim 7, wherein the per-tone values are determined after aninitialization period.
 10. The digital subscriber line transceiver ofclaim 7, wherein the time-varying noise is an impulse noise.
 11. Thedigital subscriber line transceiver of claim 7, wherein the per-tonevalues are used to adjust a minimum impulse noise protection value. 12.The digital subscriber line transceiver of claim 7, wherein the per-tonevalues are used to control trellis coding.
 13. The digital subscriberline transceiver unit of claim 7, wherein the per-tone values areprovided to the transceiver unit by a second transceiver unit incommunication with the transceiver unit.
 14. The digital subscriber linetransceiver unit of claim 7, wherein the transceiver unit uses worstcase noise during data exchange to adjust a number of bits transmittedover a tone.
 15. The digital subscriber line transceiver unit of claim7, wherein the transceiver unit uses worst case noise during dataexchange to determine whether a bit transmitted on a first tone shouldbe transmitted on a second tone.
 16. A digital subscriber linetransceiver unit, comprising: a transceiver that includes a noisemitigation engine that determines by use of a medley signal a per-tonevalue relating to a margin correction to mitigate a time-varying noisefor each tone used to transmit data bits, wherein a duration of themedley signal is varied depending on a noise impulsivity of thetime-varying noise; and a noise mitigation controller coupled to thenoise mitigation engine of the transceiver, wherein the transceiver unittransmits noise during an initialization process in order to induce abit rate limitation in a second transceiver unit in communication withthe transceiver unit.
 17. The digital subscriber line transceiver unitof claim 16, wherein the transceiver unit sets a lower limit for medleynoise based on echo noise measured after transmission of a signal havinga high crest factor.